1. Field of the Invention
The invention relates to the field of electronic circuits, and in particular, to a circuit for providing accurate current bias control for light emitting diode applications.
2. Related Art
A light emitting diode (LED) is a diode that emits photons in response to a current flow between its anode and cathode. LEDs are often used in modern lighting applications due to their durability, efficiency, and small size compared to other light sources. The range of applications for which LEDs are appropriate is continually increasing due to development of increasingly higher efficiency and higher output LEDs. For example, many types of automotive lighting elements (e.g., interior lights, external signal lights) are being updated with LED sources.
To properly power the LEDs in these high-power applications (i.e., applications in which a significant voltage difference exists between the load voltage (e.g., roughly 3.6V for a white LED) and the input supply voltage (e.g., roughly 12V for an automobile battery)), “step-down” or “buck” switching regulators are typically used. A switching regulator uses the input voltage to rapidly pulse energy into a storage element (typically an inductor), and that stored energy is then transferred into the load element (e.g., an LED). This switching methodology causes the total load current to ramp up and down between maximum and minimum current levels. A small filter capacitor at the output can be included to smooth out the current ramps to provide a constant load current into the LED. Switching regulation is therefore well-suited to driving an LED, since the light output of the LED in response to this switching behavior will be observed as a constant light output, with the actual output level of the LED being determined by the average current provided to the LED.
FIG. 1A shows a conventional step-down switching regulator circuit 100 for driving an LED D110. Circuit 100 is a buck circuit that converts a high input voltage VBATT (e.g., a 12V battery voltage) down to the desired LED drive voltage (e.g., 3.6V for a white LED) while providing a desired average drive current. Switching regulator circuit 100 includes a sense resistor R150, LED D110, an inductor L120, and a switching transistor Q140 coupled in series between a supply voltage VBATT and ground. An output capacitor C160 is coupled between supply voltage VBATT and the junction between LED D110 and inductor L120, while a Schottky diode S130 is coupled between supply voltage VBATT and the output terminal of inductor L120 (i.e., the downstream terminal of inductor L120 coupled to transistor Q140). Finally, a proportional-integral-derivative (PID) controller 101 includes inputs coupled across sense resistor R150, an input coupled to the junction between inductor L120 and Schottky diode S130, and an output coupled to the gate of switching transistor Q140.
To drive LED D110, PID controller 101 monitors the current through LED D110 by measuring the voltage drop across sense resistor R150 (which is proportional to the current through LED D110), while at the same time measuring the changing voltage at the junction between inductor L120 and Schottky diode S130. In response to the detected load (LED) current, PID generator 101 provides a pulse width modulated (PWM) control signal PWM1 to the gate of transistor Q140. Control signal PWM1 provides a square wave input signal that switches between a logic HIGH level and a logic LOW level to turn transistor Q140 on and off, respectively. Turning on and off transistor Q140 causes inductor L120 to charge and discharge to provide the desired average load current to LED D110. Meanwhile, capacitor C160 acts as a filter for this switching behavior to provide a relatively constant output voltage across LED D110.
Thus, to describe the operation of switching regulator circuit 100 in detail, when control signal PWM1 is in a logic HIGH state, transistor Q140 is turned on, and an electrical path is provided between supply voltage VBATT and ground. Current begins to flow though LED D110 and charges the magnetic field in inductor L120. As inductor L120 charges up, a current I_IND through inductor L120 (and hence, through LED D110) increases. Since supply voltage VBATT is a DC voltage, current I_IND increases linearly at a rate equal to the voltage across inductor L120 divided by the inductance of inductor L120. For example, if supply voltage VBATT is 12V, and the forward voltage of LED D110 is 3V, the voltage impressed across inductor L120 is 9V (12V−3V). Therefore, if inductor L120 has an inductance L, the rate at which current I_IND increases is 9V/L.
When control signal PWM1 switches to a logic LOW state, transistor Q140 is turned off and the voltage across inductor L120 immediately changes to a value required to maintain the level of inductor current I_IND. For example, using the above example (supply voltage VBATT=12V and LED D110 Vf=3V), the input terminal of inductor L120 (i.e., the upstream terminal of inductor L120 connected to LED D110) will be maintained at 9V. Therefore, the output terminal of inductor L120 will jump to the value of supply voltage VBATT plus the forward voltage of Schottky diode S130. If Schottky diode S130 has a forward voltage of 0.2V, the output terminal of inductor L120 immediately after switching transistor Q140 is turned off will be 12.2V (12V plus 0.2V).
Thus, immediately after transistor Q140 is turned off, inductor L120 begins discharging through Schottky diode S130 into supply voltage VBATT, thereby maintaining current flow through LED D110. However, because the current flow during this phase of the switching cycle is generated by the magnetic field stored in inductor L120, current I_IND decreases as that magnetic field dissipates. Because the voltage across inductor L120 is maintained at a relatively constant level during this discharge phase, current I_IND decreases at a linear rate that is once again equal to the voltage across inductor L120 divided by the inductance of inductor L120. For example, if VBATT is equal to 12V, and the forward voltage of LED D110 is equal to 3V, the input terminal of inductor L120 will be at 9V (12V minus 3V), while the output terminal of inductor L120 will be at 12.2V (if Schottky diode S130 has a forward voltage of 0.2V). Therefore, the voltage across inductor L120 will be 3.2V (12.2V minus 9V), and the rate at which I_IND decreases is 3.2V/L.
Conventional switching mode regulators operate either in continuous current mode (CCM) or discontinuous conduction mode (DCM). In CCM operation, inductor current I_IND cycles between two non-zero current values. FIG. 1B shows a sample graph GC of inductor current I_IND over time for CCM operation. Graph GC ramps up and down between a minimum current IC_MIN and a maximum current IC_MAX. Because of the linearly increasing and decreasing profile of graph GC, the average current IC_AVG is simply the average of maximum current IC_MAX and minimum current IC_MIN, as indicated below:IC_AVG=(IC_MAX+IC_MIN)/2  [EQ. 1]Note that this average current determination is independent of the relative slopes of the ramp up and ramp down portions of the waveform for inductor current I_IND.
During DCM operation, the inductor current is allowed to fall to zero for a portion of the discharge cycle. In other words, the magnetic field in the inductor is allowed to collapse, so that current no longer flows through inductor L120 (and hence LED D110). After a period of time, control signal PWM1 turns transistor Q140 back on, and current I_IND begins increasing from zero. FIG. 1C shows a sample graph GD of inductor current I_IND over time for this DCM operation. Graph GD initially ramps from zero to a maximum current ID_MAX, and then ramps back down to zero, remaining at zero for an offtime duration D. The average current ID_AVG for DCM operation is therefore equal to half of the maximum current ID_MAX scaled by the proportion of time inductor current I_IND is at a non-zero value, as indicated below:ID_AVG=(ID_MAX/2)*(1−D/T)  [EQ. 2]where T is the period of the current waveform (i.e., the time between successive peaks).
As noted above, the output of an LED is determined by the average current supplied to the LED. Therefore, the accurate generation of average current IC_AVG during CCM operation and the accurate generation of average current ID_AVG during CCM operation are important for proper LED function. Unfortunately, accurate average current control for either CCM or DCM operation can be extremely complicated. For example, when switching regulator circuit 100 (in FIG. 1A) is operating in CCM mode, the values of maximum current IC_MAX and minimum current IC_MIN are determined by the duty cycle of control signal PWM1. Specifically, the logic HIGH portion of each cycle of control signal PWM1 must be long enough for inductor current I_IND to ramp from minimum current IC_MIN to maximum current IC_MAX, while the logic LOW portion of each cycle must be long enough for inductor current I_IND to ramp down from current IC_MAX to current IC_MIN. However, due to variations in operational characteristics (e.g., the actual value of supply voltage VBATT, the actual forward voltage of LED D110, and the actual inductance of inductor L120 will all vary to some degree from circuit to circuit), additional circuitry must be used to measure the actual value of inductor current I_IND generated in response to the switching control. Furthermore, the feedback loop resulting from such additional current monitoring circuitry can require sophisticated control to properly regulate the resulting control signal PWM1. Typically, a PID controller (e.g., PID controller 100) is used, which further increases implementation complexity and cost. Similar drawbacks apply to the use of DCM mode, with even greater difficulties due to the addition of the off-time period during each cycle (i.e., offtime duration D in FIG. 1C).
Another issue for conventional switching regulator circuits (such as circuit 100) is that monitoring the load current to allow proper functioning of a PID controller requires that a sense resistor be placed in-line with the LED. The sense resistor must be relatively large to minimize unnecessary power consumption, and is therefore typically external to the switching regulator circuit. However, this external placement then mandates that the packaging for the switching regulator circuit include additional pins to enable measurement of the voltage across the sense resistor. The resulting increase in pin count can preclude the use of smaller, more desirable chip packaging for conventional switching regulator ICs.
Accordingly, it is desirable to provide a simple switching regulator that can be easily configured to provide an accurate average load current.